The band-gap voltage reference circuit is well-known to the designers of analog circuits. This reference circuit provides a constant voltage as independent as possible of the environmental temperature at which the circuit operates. This type of circuit is present in many systems manufactured with integrated circuits. For example, a constant voltage reference is required for analog/digital converters. Such converters compare the value of a voltage reference signal against the value of samples to be converted.
The output voltage of the voltage reference circuit can be temporarily connected to various users. The circuits of such users constitute a load which the voltage reference circuit must be able to drive. When one or more of these load circuits require the use of the voltage reference output, the voltage reference signal can undergo a voltage degradation due to insertion of a load, for a certain period of time.
Voltage degradation under these circumstances occurs for only a certain period of time. The voltage reference circuit, however, must have the capability to quickly react to this degradation and rapidly restore the output voltage to the correct reference voltage value. Errors may occur if a load circuit is added to the voltage reference output during this degradation period. Thus, quick restoration to the correct voltage value is necessary so that the voltage reference circuit is available to user circuits without excessive delay. Additionally, a quick restoration period is necessary in order to minimize the possibility of errors due to the addition of load circuits during the degradation period.
Numerous circuit arrangements are known which provide a band-gap voltage reference of the time-continuous type (i.e., of the type which can be used for 100% of the time). One such circuit is disclosed in Gray and Mayer, "Analysis and Design of Analog Integrated Circuits", published by John Wiley & Sons. These circuits are generally manufactured using bipolar junction technology because their operating principle is based on intrinsic properties of bipolar junction transistors (BJT) as described in the above noted publication.
Complex systems are often executed in a single CMOS integrated circuit. Current technology enables an integrated circuit to be fabricated with both analog and digital circuits in a compact manner on the same integrated circuit. As is known in the art, such integrated circuits can contain vertical bipolar parasitic transistors which are operable at low frequencies. Such parasitic transistors can be placed in the integrated circuit without additional steps being added to the manufacturing process. Thus, it is currently possible to produce a band-gap voltage reference circuit having just two bipolar junction transistors, and the remainder of the integrated circuit fabricated in MOSFET technology (see Gray and Meyer above).
The operating principle of a band-gap voltage reference circuit is based on the compensating increases and decreases in the rate of voltage change due to changes in environmental temperature. That is, the voltage between the base and the emitter of one bipolar transistor decreases with the environmental temperature at the rate of approximately 2 mV/.degree. C. Conversely, the proportional difference in the base-emitter voltage between two bipolar transistors operating at different current intensities is governed by the equation kT/q. The equation kT/q is given where k is Boltzmann's constant, q is the charge of the electron, and T is the temperature in degrees Kelvin. and C3. Capacitors C1, C2 and C3 are selectively switched utilizing switches S1, S2, S3 and S4. The switches are transitioned synchronously at appropriate frequencies so as to obtain a voltage reference signal V.sub.ref which is the weighted sum of V.sub.be and .delta.V.sub.be at the output of the operational amplifier OP. The voltage reference V.sub.ref in this configuration is represented by the following equation: EQU V.sub.ref =a.sub.1 V.sub.be +a.sub.2 .delta.V.sub.be
This equation is given were the weights a.sub.1 and a.sub.2 are respectively equal to C1/C3 and C2/C3. Substituting the weights in the above equations results in the following equation: ##EQU1## Weights a.sub.1 and a.sub.2 (that is, the values C1, C2 and C3) can be chosen so as to make the voltage reference signal V.sub.ref independent of the environmental temperature. Thus, the values of the capacitors C1, C2 and C3 are selected to make the voltage reference signal V.sub.ref independent of temperature changes such that any increase in the voltage drop across one transistor due to a change in environmental temperature will be offset by a corresponding decrease in the voltage drop across the other transistor due to the same change in environmental temperature; and, vice versa.
The switches control the capacitors, but the switches also cancel the offset of the operational amplifier. An initial reset step is accomplished by short-circuiting the input and the output of the operational amplifier through switch S3 (closed or ON). Switches S1 and S2 are in a Utilizing the equation kT/q, we find the proportional difference in the base-emitter voltages of two bipolar transistors operating at different current levels increases with the temperature at the rate of approximately 0.2 mV/.degree. C. Thus, the base-emitter voltage of one transistor will decrease at a rate of 2 mV/.degree. C., while the difference between the base-emitter voltages of two transistors operating at different currents will increase at a rate of 0.2 mV/.degree. C.
A substantially temperature-independent voltage is obtained by multiplying one voltage difference by an appropriate factor and adding it to the other voltage difference. That is, a decrease of the voltage difference V.sub.be in the first bipolar transistor due to a temperature change is compensated by the increase of the voltage difference .delta.V.sub.be of the two bipolar transistors operating at different current intensities due to an identical temperature change; and, vice versa. Thus, the temperature independence of this circuit is based on the corresponding rates of change in the voltage difference of the first transistor and the corresponding rate of change in the voltage difference of the two transistors operating at different current values.
A particular type of band-gap voltage reference circuit is the sampled-time reference disclosed in Vittoz, "Design of High Performance Analog Circuits on Digital CMOS Chips", JSSC, June 1985. A known circuit of this type is shown in FIG. 1, wherein two constant current sources I.sub.1 and I.sub.2 of unequal value drive the emitters of respective diode-connected bipolar transistors T1 and T2. The voltages across transistors T1 and T2 are applied to a switched-capacitor integrator.
The switched-capacitor integrator comprises an operational amplifier OP combined with capacitors C1, C2 position such that the input voltage is derived from a first common node of a first transistor operating at a first current value.
In a second step, switch S3 opens and the output of the operational amplifier is fed back to the operational amplifier OP through capacitor C3. The switches S1 and S2 are also moved to a position where the input voltage is derived from a second common node of a second transistor operating at a second current valve. The reference voltage is generated during this second step and remains available until a subsequent reset step. Usually, this reference voltage availability step is much longer than the reset step for practical reasons.
The reference circuit described above has the disadvantage of being slow in restoring the output voltage value to the correct reference voltage value when it is loaded by several user circuits in succession. The interference of capacitors C1 and C2 is a primary reason for the slow speed at which the reference circuit restores to the correct voltage reference value after loading.
In order to accurately compensate for temperature changes, there must be a rather large ratio between the capacitance of capacitors C1+C2 and the capacitance of capacitor C3. This ratio between C1+C2 and C3 must usually be higher than 10. After the reset step, any feedback voltage of the operational amplifier from the output to the inverting input passes through capacitor C3. This feedback voltage is attenuated by capacitors C1 and C2 largely due to the high capacitive ratio.
This capacitive attenuation of the feedback signal causes the operational amplifier to operate within a reduced bandwidth. This attenuation prevents maximum utilization of the operational amplifier's potential bandwidth. The higher the ratio between C1+C2 and C3, the narrower the bandwidth. The settling time of an operational amplifier depends on its passband or the bandwidth utilized by the operational amplifier. A reduced passband translates into a slower regeneration period for the voltage reference signal V.sub.ref. Thus, the attenuation caused by capacitors C1 and C2 narrows the reference circuit passband which slows the regeneration period of the circuit.
In order to shorten the regeneration period, a faster than necessary operational amplifier can be used. Faster operational amplifiers, however, occupy a much larger area of the integrated circuit and absorb more power than the smaller operational amplifiers. Integrated circuit area and power consumption should be minimized under most circumstances. Thus, in many instances, these faster operational amplifiers cannot be effectively utilized in an integrated circuit with area constraints and power consumption limitations. Additionally, in many instances the slower operational amplifiers cannot be utilized in the voltage reference circuit because of the excessive delay times.